Synchronous detection, long used in telecommunications because of performance potential, can now be effectively applied to sensor interface circuitry due to advances in low-cost ICs. This circuit (see the figure) employs a synchronous-detection scheme to measure the resistance of RTDs (resistance-temperature detectors) with self-heating errors of less than 0.001°C.
Conventional circuits require a large current through the sensor so that the smallest temperature change to be measured results in a voltage change larger than the total system noise, drift, and offset. For example, a 0.3-Ω/°C RTD with a 0.1% system resolution requirement and a system offset and drift of only 0.3 mV needs an excitation current in excess of 10 mA. The power dissipated in the RTD causes its temperature to rise above its ambient by:
ΔT = (I2RRTD) θ
where ΔT = the change in temperature due to internal power dissipation; I = excitation current; RRTD = the value of the RTD in ohms; and θ = the self-heating effect in °C/mW.
In the previous example, upon power on, an RTD with θ = 0.05°C/mW, and a nominal value of 100 Ω at 0°C using a 10-mA excitation current will have a drift error of 0.5°C. That's due to the self-heating effect, which may take several minutes to settle (see the table).
Waiting for the system to settle is pointless, because a self-heating-induced gain error remains. Drift errors can also be caused when the medium measured changes flow rate. This will cause a variation in the chill effect, which in turn will change ΔT.
Reducing the excitation current to 100 µA will, of course, effectively lower this self-heating error by 10,000. However, the transducer output will be reduced by 100 (to 0.385 µV per 0.01°C, for an RTD with α = 0.385 Ω/°C). (α is a measure of an RTD's resistance slope.)
That's well below the offset, noise, and drift of amplifying elements, effectively rendering the transducer output invisible.
In the synchronous technique, the RTD is excited by a 1-kHz ac waveform. The RTD's varying resistance then modulates this sine-wave carrier. The modulated waveform, which conveys information about the RTD value and hence sensed temperature, is demodulated using the same ac waveform as its reference. Because the demodulation process employs the same reference, uncorrelated perturbations like noise, offset, and drift can be distinguished and filtered out from variations caused by RTD modulation. In this application, the frequency stability and distortion of the nominal 1-kHz carrier aren't critical, because the same waveform is used for modulation and demodulation.
The circuit in the figure uses a 100-µA peak alternating current through the RTD to create an alternating voltage across it that's proportional to temperature. IC1b (half of an AD706 precision op amp) supplies a reference voltage so that at some reference temperature, RT1 may be trimmed to yield a zero differential input voltage to the instrumentation amplifier, IC2 (an AD620). IC2 amplifies this differential voltage and RG sets the gain of the instrumentation amplifier to 408. The output of IC2 is then demodulated with a synchronous detector IC, IC3. IC3's output is low-pass filtered, removing all uncorrelated disturbances, such as noise, offset, and drift, while retaining a dc voltage that's proportional to the change in resistance of the RTD from its nominal value. IC4 (a precision operational amplifier) provides a noninverting gain of 10. Potentiometer RT2 supplies the trim adjustment for gain accuracy.
The relationship between the RTD's resistance and the output voltage is then:
Vout = 2/ π Vpk/10 kΩ × ΔRRTD × G
where ΔRRTD = the change in RTD resistance from nominal value; and G = total system gain.
For an RTD with α = 0.385 Ω/°C, the output scale then becomes 100 mV/°C of deviation from the reference temperature.
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