Designers of networking equipment often request a model of the transformer and common-mode-choke module used in 10-/100Base-T applications. With that model, they can simulate the module's performance with the associated driver IC and other related components. Before building physical prototypes, the circuit can then be optimized. This guarantees that the final design will comply with the requirements of the chosen networking protocol.
In some cases, simulation results lead to changes in the design of the driver IC. But more often, they request that the magnetics designer modify the transformer and/or choke module to achieve the desired performance.
A typical model, which can be used for PSpice analysis, consists of two sections (Fig. 1). When Pulse customers request this model, it's supplied with parameters based on measured values, rather than the specified limits of the device. The values shown in the diagram are for a typical transformer module.
To properly apply this model, designers should understand how its various parts are derived. They also need to know how those parts affect the output waveform and the EMI noise entering the transformer. Although the model treats the transformer and common-mode choke separately, those two sections do interact. When they're combined, the parasitic elements of the common-mode choke enhance the transformer's performance.
The source impedances (RA and RB) represent the input impedance from the driver device. They're equal for each half of the differential-circuit operation. To be a critically damped circuit, the sum of RA and RB must equal the reflected load impedance. If they aren't equal, the return loss of the transformer will be affected.
The primary winding resistances (RP1 and RP2) are calculated by: wire resistance per inch × number of turns × winding length per turn around the core. This parameter is minimal. It affects the amount of insertion loss within the passband frequencies.
The leakage inductances of the primary (LLP1 and LLP2) and secondary (LLS1 and LLS2) are the winding-wire inductances for each half of both windings. For the best common-mode rejection and EMI, every winding half must be balanced with its other half.
When designing 10/100Base-T trans-former and common-mode-choke magnetics, the ultimate goal is to minimize leakage. This parameter drastically affects return loss at the higher frequencies (Fig. 2).
Low leakage inductance produces a return loss with a resonant dip. This dip allows the return loss to be greater at the higher frequencies. As the leakage inductance increases, that return loss becomes the rounded curve. It then has less return loss at the same frequency.
A shunt resistance (RCX) stands for the parallel-equivalent core loss across the transformer's primary side. This resistance, which represents the eddy-current losses in the core, increases with frequency. It has the greatest effect on the midband insertion loss and the pulse-backswing-voltage amplitude, better known as the flyback voltage. This voltage varies by the square of the number of turns on the core.
The distributed capacitances (CDP and CDS) are the turn-to-turn capacitances between wires of the same winding. When the secondary one, CDS, is reflected back to the primary side, that side multiplies its value by the inverse square of the turns ratio. CDS then becomes parallel with the primary distributed capacitance, CDP. This total capacitance is seen as a shunt capacitance by the driver chip.
Increasing distributed capacitance can improve the return loss of the transformer up to a point of diminishing returns. Exceeding this value will produce a deterioration of that loss (Fig. 3).
Lying between the primary and secondary windings of the transformer is the interwinding or coupling capacitance (CWX1 and CWX2). Such coupling capacitance impacts the high-frequency roll-off of the transformer. The higher the capacitance is, the lower the cutoff frequency will become. This parameter will affect return loss as well (Fig. 4).
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