[Design Application]
Design Engineers Battle The Dark Side Of Electromagnetism
An arsenal of bypass and decoupling strategies help fight the interference caused by increased pc-board densities, switching speeds, and switched power.
Today's higher population densities on pc boards, faster switching speeds, and greater switched power contribute to the increased probability of interference between circuits, modules, and systems. This means emissions requirements are much tighter. Design engineers now recognize the need for more rational and effective bypass and decoupling strategies as they combat the dark side of electromagnetism.
The old "rabbit's foot" approach of sprinkling capacitors around the board proves much less effective under the new conditions, possibly even aggravating an existing problem. All capacitors, leads, and board traces have associated series inductance which, at the higher switching rates, is neglected at one's peril. These formerly negligible reactances can interact with the capacitance to produce not only zeroes, but also poles that could yield unpleasant or mystifying results.
We approached this issue in a previous article ("Tuned Decoupling Tames Noise In Switching Circuits," Electronic Design, July 6, 1998, p. 42) by exploring the use of capacitor-based decoupling networks tuned via pc-board trace inductance to specific frequencies that we desired to suppress. The article detailed the advantages of the technique as well as the caveats of employing it. Here, we revisit tuned decouplers and look at the use of discrete inductors to improve and extend the decouplers' effectiveness.
Effective decoupling requires that the decoupler provide the ac content of the switched current while the main power source supplies the average current, and therefore, all of the energy. In this way, high-frequency currents stay off of the power and ground buses and remain confined to small (decoupling) loops near the target, thereby minimizing emissions. Therefore, the primary analytical tool is a model based on an ac-current-source representation of the switching component, or network (Fig. 1). In this model, iO is the ac current through the switched unit, iS is the part of iO that flows through the inductance LS of the power-bus trace, and Z is the impedance of the decoupling network. The response iS(f)/iO(f) is obtained from this model.
An empirical comparison for the analytical inferences was possible with a hardware implementation (Fig. 2). Top-side (component) traces were sized for desired inductance value, according to the formulas of Rostek ("Avoid Wiring-Inductance Problems," electronic design, Dec. 6, 1974, p. 62). The entire bottom side of the board was used for the ground plane.
The power bus was connected to a 10-V dc power source and was bulk-decoupled by two 47-µF tantalum capacitors at the connection point. The 2N7000 FET switch was driven by a 5-V square wave at a 1-MHz rate with a 50% duty cycle. A magnetic field probe monitored radiation off of the power-bus trace, and the obtained information was displayed on a Hewlett-Packard spectrum analyzer.
For additional insight, a Spice model of the hardware was used to study the various decoupling configurations. In all Spice runs, 100 cycles of the driving signal were sufficient to attain steady-state values. Three circuit representations were studied, and in general, the three stood in agreement.
The three basic decoupling networks included a trace-tuned, capacitor-based parallel branch or branches (Fig. 3); a pi filter with series discrete inductor and trace-tuned, capacitor-based parallel branches (Fig. 4); and an LC filter with discrete series inductor and trace-tuned, capacitor-based parallel branches (Fig. 5).
The capacitors C1 and C2, shown schematically in these figures, were MuRata Erie radial-leaded ceramic devices, made with X7R dielectric. The reactances represented by L1 and L2 are trace and/or lead inductances. The reactance represented by LX was implemented with Coilcraft 90-13 axial-leaded inductors.
Any 1-µF ceramic capacitor has an inherent antiresonance (i.e., minimum impedance) at 5 MHz, indicating an intrinsic inductance of about 1 nH. Adding 24 nH of series trace/lead inductance yields the antiresonance at about 1 MHz. Similar considerations hold for a 0.033-µF capacitor, which has an inherent antiresonance at 27.5 MHz.
For convenience, the investigation assumed that the spectral region of concern extended from the fundamental (1 MHz) to the fifth harmonic (5 MHz). You can find the empirical results for important points along the decoupler development path in Figures 6, 7, 8, 9 and 10. The pole frequencies were computed from the ac model. Additional plots of hardware results can be accessed on the Web at www.elecdesign.com.
Tuned decoupling networks can very effectively suppress emissions produced by switching circuits. A single-branch tuned decoupler acts to quench emissions by producing a zero at the frequency to which it's tuned. Additionally, it can reduce emissions at higher frequencies. Compare Figure 7 and Figure 11 with Figure 6 (the no-decoupling case). When a single frequency dominates the emission pattern, a single-branch decoupler tuned to the offending frequency obtains a good quench.
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