[Design View / Design Solution]
Improve EMC In Class D Amplifier Applications
Besides reducing EMI, new modulation techniques and filter architectures provide the cost benefits and audio performance of class AB designs.
NEW MODULATION TECHNIQUES Due to the rising interest in class D amplifiers, several manufacturers have introduced modulation schemes that provide independent control of the two halves of the H-bridge. These schemes offer two key advantages:
For very low audio signals and at idle, virtually no differential switching is present across the load. This improves quiescent-current consumption over PWM designs.
The minimal-pulse CM switching helps to reduce the turn-on and turn-off transients. The dc idle level (post filtering) on each of the BTL "legs" is close to GND, which minimizes any mismatch from filter components or stray capacitance (that can give rise to audible click-and-pop when the amp is enabled or disabled).
There clearly are some advantages to the new techniques, but the amplifier outputs no longer are mirror images of each other (Fig. 2). The waveforms shown (representing the MAX9704 stereo class D amplifier) have a high level of CM content. Output-filter requirements are, therefore, different from those of an amplifier with the traditional differential and complementary PWM outputs. Compared with PWM, the new modulation scheme includes a high level of CM signal, and any output-filter design should take that into account. A traditional differential filter topology may provide poor results, as the following example shows.
Figure 3a depicts a traditional LC, PWM class D output filter implemented with ideal values. For simplification, the speaker load is represented as a pure 8-Ω resistance, and the inductor's dc resistance is assumed to be negligible. Some straightforward Spice simulations can highlight the problem.
Figure 3b shows the response of the Figure 3a filter driven by a differential input signal. Each output node (FILT1, FILT2) is plotted with respect to GND. The values chosen create a second-order slope above 30 kHz and a well-controlled transition. Group delay is flat across the audio band at approximately 4 µs.
Figure 3c shows the same filter output driven with a common-mode signal. Again, each output is plotted with respect to GND. The result (note the shifted Y axis!) is heavily peaked and obviously very underdamped. That's easily understood, considering how the filter appears to a CM signal. Because the simulation provides ideally matched inductors and capacitors, the differential signal across the resistive load is zero. Subsequently, it has no damping effect on the LC components.
L1 interacts with C1 (as does L2 with C3) to provide the peaked response. In the time domain, this condition would indicate heavy overshoot and ringing. C2 contributes zero when driven common-mode, meaning the filter's cutoff frequency (or more accurately in this case, its resonant frequency) is higher than that of the differential case.
It probably isn't a problem if the output spectrum has zero CM energy at that frequency. If the peaking frequency coincides with the class D switching frequency, large voltage-output excursions can appear at the speaker and the cabling.
Further, the MAX9704 in its spread-spectrum mode (SSM) may excite the underdamped filter by producing appreciable noise energy above the audio band. SSM is a pin-selectable option in which the high-frequency switching energy is "whitened" and lowered in amplitude by randomizing the switching period on a cycle-to-cycle basis, also easing EMI compliance in a filter-less design.
POSSIBLE SOLUTIONS One solution is to preserve the basic architecture of Figure 3a, but add damping elements that suppress the highly resonant common modes. Take a look at the addition of two series RC elements from each output node to GND in Figure 4a. If efficiency isn't important, you can simply add resistors to GND. But capacitors C4 and C5 help to minimize excessive power dissipation in R1 and R2.
The values of C4 and C5 impose a tradeoff. They must be large enough to allow R1 and R2 to damp out the peaking, but small enough to minimize the power loss at high audio frequencies (usually up to 20 kHz). This tradeoff is made easier if the CM cutoff frequency is much higher than the differential-mode frequency, a condition implemented by increasing the ratio of C2 to C1 and C3.
By increasing the CM cutoff frequency, C4 and C5 can be made smaller and R1 and R2 larger, minimizing the audio-frequency power loss into R1 and R2. Pushing the CM cutoff frequency too high, however, allows more common-mode on the cables, so you must determine a reasonable limit in the ratio between the differential and CM –3-dB points. For this filter, we've adopted 1:5 for that ratio.
Figure 4b shows the filter of Figure 4a driven differentially, and Figure 4c shows the response when driven commonmode. Note the higher-frequency, CM cutoff in Figure 4c (–3 dB at around 110 kHz, versus 28 kHz for the differential case), with gentle but well-controlled peaking. This cutoff is well above the highest audio frequencies (and below the class D switching-frequency fundamental), so it should be of little consequence.
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