[Design View / Design Solution]
Zero-Drift IA Takes The Strain Out Of Sensor Measurements
Instrumentation amplifiers can abet many sensor applications, from ratiometric bridges to low-side current sensing.
Even if VOS can be calibrated out during manufacturing, the drift of input offset voltage (with temperature and time) can be of greater concern than the initial dc offset itself. Such drift errors are best tackled through the use of active circuitry within the chip.
Perhaps the single most important source of ac error is noise, which is inherent in the semiconductor chip design and process. Because most sensor signals are amplified by highgain blocks, the input referred noise is also amplified by that same gain. Noise comes in two forms: pink noise (also called 1/f or flicker noise) and white noise. Pink noise is more critical at lower frequencies (<100 Hz or so), and white noise generally determines the chip performance at higher signal bandwidths.
In traditional low-noise analog-circuit design, bipolar transistors are often preferred for use in input-stage circuitry, especially if low levels of pink noise must be achieved. Pink noise originates as recombination effects at defect sites on the semiconductor surface. Therefore, the noise of CMOS devices tends to demonstrate a larger magnitude and a higher corner frequency than the noise developed by bipolar devices. (The frequency at which pink noise density equals white noise density is defined as the noise corner frequency).
Most sensors prefer high-impedance inputs, which forces the use of CMOS front ends on IAs. This, in turn, would seem to force one to live with the accompanying higher levels of lowfrequency noise. Fortunately, zero-drift circuit-design techniques that continuously cancel out input offset voltages also tend to cancel the low-frequency input pink noise.
COOL NEW ARCHITECTURES ARE REALLY HOT A traditional IA uses three op amps to create an input buffer stage and an output stage (Fig. 1). The input buffer stage provides all differential gain, unity common- mode gain, and a high-impedance input. The differential amplifier output stage then provides a unity differential gain with zero common-mode gain. This IA works quite well in many applications, but its simplicity hides two significant drawbacks: the usable input commonmode voltage range is limited, and its ac CMRR is limited.
IAs based on three-op-amp architectures suffer a restrictive transfer characteristic (Fig. 2). Their architecture can allow the outputs of buffer amplifiers A1 and A2 to saturate into the power-supply rails during a certain combination of input common-mode and input differential voltage. In this condition, the IA no longer rejects input common-mode voltages.
As a result, the data sheet for most three-op-amp IAs shows a plot of the usable input common-mode voltage versus output voltage. Because output voltage is simply a scaled version of the input differential voltage, the two axes of this plot could also be labeled “input common-mode voltage versus input differential voltage.” The gray area within the hexagon depicts the “valid” zone of operation, where the outputs of amplifiers A1 and A2 aren’t saturated into the power-supply rails.
Note that the graph of Figure 2 has an important implication for single-supply applications. Common-mode voltages can easily approach the circuit ground, to which the gray zone doesn’t extend! Certain applications (such as low-side current sensing) can’t use a traditional three-op-amp IA, because the input common-mode voltage equals the ground potential.
Three-op-amp IAs achieve high common- mode rejection at dc by matching on-chip resistors around the differential amplifier, but the feedback architecture of such IAs can substantially degrade the ac CMRR. To overcome this and other drawbacks, alternate IA architectures have been developed. The 2-gm indirect current-feedback approach, for instance, has found considerable success (Fig. 3).
This architecture consists of two matched transconductance amplifiers and a high-gain amplifier. Because the matched amplifiers have the same gm, they develop equal differential voltages at their inputs, and the output voltage is therefore determined by the resistor divider ratio Rf/Rg. The output commonmode voltage is set by the voltage at the REF pin. Voltage-to-current conversion implemented by the input gm amplifier inherently rejects the input commonmode voltage, giving the amplifier a high dc and ac CMRR.
The indirect current-feedback IA architecture allows a full output-voltage swing even when the input commonmode voltage equals the negative supply rail. Thus, it offers an expanded range of operation not obtainable with the threeop- amp IA architectures. Examples of this IA type from Maxim Integrated Products include the MAX4460/1/2 and the MAX4208/9.
OFFSET-CANCELLATION TECHNIQUES: CATCH THE DRIFT? As mentioned above, two important specifications for IAs are pink noise (also called 1/f or flicker noise) and input offset voltage and its drift (versus temperature and time). Because 1/f noise is a low-frequency phenomenon, many of the circuit techniques used to achieve “zero drift” and cancellation of input-offset voltage also remove 1/f noise. These techniques include sampling amplifiers, auto-zeroing amplifiers, chopper amplifiers, chopper-stabilized amplifiers, and chopper-chopper-stabilized amplifiers (e.g., the MAX4208).
Sampling techniques based on flying capacitors have also been applied to IAs for the purpose of auto-correcting input offset voltages. However, since a sampled input isn’t a true high-impedance structure, system-level accuracy can be compromised by a mismatch in the source resistances (such as those found in certain unbalanced bridges).
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