GaN Power Transistors: Master Strokes on a Power-Supply Canvas

2. Wide bandgap semiconductors such as GaN and SiC offer superior performance over silicon in power applications.

Higher switching frequencies reduce power converter size, weight, and cost because they allow the use of smaller capacitors, inductors, and transformers. Switching to GaN power devices can cut the size of a power supply by up to 50%.

Let’s examine in more detail the advantages of switching to GaN, beginning by reviewing the simplified high-frequency model of a power MOSFET in Figure 3. This model applies to both silicon and GaN technologies.

3. A simplified high-frequency equivalent circuit of a MOSFET. (Source: Author)

It includes parasitic junction capacitances C_GD, C_GS, and C_DS. In the datasheet, these are stated as the input capacitance (C_ISS), the output capacitance (C_OSS), and the reverse transfer capacitance (C_RSS), where:

C_RSS = C_GD

C_ISS = C_GS + C_GD

C_OSS = C_DS + C_GD

The parasitic junction capacitances are formed from a series combination of a bias-independent oxide capacitance and a bias-dependent depletion capacitance. C_OSS decreases for lower values of R_DS(ON). For a given device, C_OSS is also nonlinear and decreases significantly above a V_DS threshold.

The shape of the curve is significantly different for Si and GaN devices. As shown in Figure 4, for a GaN FET, C_OSS is proportional to R_DS(ON) and varies very little for V_DS < 30 V.  

4. Variation of C_OSS with V_DS and R_DS(ON) for GaN and Si FETs. (Source: TI: “Comparison of GaN- and Silicon FET-Based Active Clamp Flyback Converters” PDF, p. 6)

For a silicon FET, on the other hand, C_OSS steadily increases as V_DS decreases below 30 V, but the increase isn’t proportional to R_DS(ON). For the device in Figure 4 with R_DS(ON) = 680 mΩ, C_OSS at V_DS = 30 V is about 100X larger than its value at V_DS = 400 V. For another device with R_DS(ON) = 180 mΩ, C_OSS is almost 300X larger over the same V_DS range.

A silicon MOSFET also has a body diode—it’s an intrinsic PN junction formed during the manufacturing process. One consequence of the body diode is an undesired reverse-recovery charge when the diode turns off. GaN FETs don’t have an intrinsic body diode, so there’s zero reverse-recovery charge.

Application Example: The Active Flyback Converter

As we can see, the characteristics of Si and GaN devices are significantly different, but how does changing from silicon to GaN improve performance in a power supply? Let’s use the active-clamp flyback (ACF) converter as an example. The ACF is a popular choice for small power adapters because it combines high power density with high efficiency.

Figure 5 shows the ACF circuit. The active clamp consists of a high-side secondary switch (Q_H) in series with a clamp capacitor (C_CLAMP). This circuit uses silicon devices, hence the body diodes.

5. The active-clamp flyback converter, showing the MOSFET body diodes and some of the equivalent circuits.

Also shown in Fig. 5 are some essential equivalent circuit elements: the output capacitances (C_OSS) for Q_H and Q_L; and the secondary rectifier capacitance (C_SEC). The transformer can be modeled as a transformer leakage inductance (L_LK) in series with a primary-side magnetizing inductance (L_PM).

The purpose of the active clamp is to recycle the energy stored in the transformer leakage by storing it in C_CLAMP. The circuit then delivers the energy to the output later in the switching cycle to minimize the voltage spike at the transformer primary side, thus reducing the voltage stress on the main switch Q_L.

Controlling the timing of the active clamp switch can also eliminate switching loss by turning on Q_L at the zero-volt switching point (ZVS operation). ZVS operation allows the ACF switching frequency to be higher, reducing the size of the power supply.

Figure 6 shows the switching events in the ACF sequence, beginning with primary power switch Q_L and clamp switch Q_H both off. Note that both I_CLAMP and I_M are bidirectional as shown in Fig. 5 above: the direction is denoted in the discussion by, for example, I_M(+) and I_M(-).

6. The principal currents and voltages of the ACF during a ZVS switching cycle. (Source: TI: “Comparison of GaN- and Silicon FET-Based Active Clamp Flyback Converters” PDF, p. 2)

A summary of the active-clamp ZVS operation follows. Consult this paper for an in-depth discussion.

1. Q_L turns on: I_M(+) is linearly increasing, storing energy in the primary-side magnetizing inductance L_PM.
2. Q_L turns off: I_M(+) charges the junction capacitance of Q_L (C_OSS(QL)) and discharges both the junction capacitance of Q_H, (C_OSS(QH)) and the junction capacitance of the secondary rectifier (C_SEC). Therefore, the current on Q_L (I_QL) decreases, the clamp current (I_CLAMP) increases, and the secondary rectifier current (I_S) increases with V_SW rising from 0 V to a high level. I_M(+) flows through the body diode of Q_H to charge C_CLAMP.
3. Q_H turns on: Q_H is conducting, NVOUT starts to demagnetize L_PM, so I_M(+) starts to decay, and L_PM releases its energy to the output. At the same time, C_CLAMP absorbs the L_LK energy by resonating with L_LK; therefore, I_CLAMP is in the positive direction.
4. Q_H turns off: The negative magnetizing current, I_M(-), starts to discharge C_OSS(QL), charge C_OSS(QH), and charge the C_SEC, so V_SW decays to 0 V.
5. The sequence then repeats: Q_L turns on as V_SW reaches 0 V, giving ZVS operation.

For ZVS to occur, the energy stored by L_PM with current I_M(-) must be greater than or equal to the energy stored in the lumped capacitance (C_SW):

The equation also shows that a larger C_SW requires higher I_M(-) for ZVS.

The ACF topology achieves higher overall efficiency than other flyback designs. Figure 7 depicts a qualitative comparison. The ACF’s clamp and switching losses are virtually zero, but improvements can still be made in other areas.

7. A comparison of the relative losses in three flyback converter designs. (Source: TI: “Comparison of GaN- and Silicon FET-Based Active Clamp Flyback Converters” PDF, p. 3)

Building up the additional negative current increases the flux density. Thus, the ACF core loss is slightly higher than that of a flyback design with a passive clamp. In addition, the ACF clamp current flows in the transformer primary winding during demagnetization time, increasing the winding loss. if the transformer losses become too large, they can negate the efficiency gains of the ACF.

Switching to GaN Reduces ACF Power Losses for Increased Efficiency

Changing to a GaN FET reduces these losses. GaN’s lower C_OSS results in lower peak-to-peak I_M. A lower I_M, in turn, leads to lower core loss; lower primary current for less winding loss; and lower clamp current for less conduction loss in Q_H.

Figure 8 compares the primary-current and clamp-current waveforms of two ACFs with Si and GaN FETs, respectively. Each FET has a similar R_DS(ON), and each circuit uses a Schottky diode as the secondary rectifier. The GaN device gives a reduction of more than 22% in peak-to-peak I_M and RMS primary current I_PRI compared to a Si FET.

8. A comparison of current waveforms highlights GaN’s superior ACF performance. (Source: TI: “Comparison of GaN- and Silicon FET-Based Active Clamp Flyback Converters” PDF, p. 8)

GaN Improves ACF Efficiency at Light Loads

It’s not only the difference in magnitude of C_OSS between Si and GaN that affects the ACF performance. The nonlinearity difference has an effect, too, particularly in the efficiency of the ACF under light loads.

As the load current becomes lighter, the current-mode control loop reduces the positive peak current I_M(+) to regulate the output power, while the negative magnetizing current I_M(-) doesn’t change, assuming a constant input voltage. I_M(+) delivers energy to the output, while I_M(-) stores the circulating energy for ZVS operation.

If I_M(+) and I_M(-) become closer in magnitude as the load decreases, the transformer’s efficiency drops, and the losses increase since the contribution of the circulating energy is greater.

A silicon-based ACF with its highly-nonlinear C_OSS has a larger I_M(-), especially at high input voltage V_BULK. At light load, the impact becomes more significant, and the efficiency deteriorates very quickly. The measurement result in Figure 9 shows that the efficiency difference between 50% load and 25% can be as high as 7.3%. 

9. The efficiency of an ACF with a silicon FET degrades more under light loads than a GaN-based design. (Source: TI: “Comparison of GaN- and Silicon FET-Based Active Clamp Flyback Converters” PDF, p. 11)

In comparison, a GaN’s lower C_OSS results in a relatively lower I_M(-). Its impact is smaller under light loads, though, with only a 2.6% difference in efficiency between 50% load and 25% load.

Driving the GaN: The Other Piece of the Puzzle

The GaN transistor provides significant performance improvements to the ACF and other power designs, but it presents challenges for the driver circuit. Designers can’t just swap in a GaN device for its Si counterpart. For one thing, traditional GaN FETs are depletion-mode (normally-on), whereas Si MOSFETs are normally-off enhancement-mode devices.  

To provide drop-in replacements for Si MOSFETs, GaN FET switch suppliers must use another switch in series to provide the required normally-off functionality. A typical solution pairs a depletion-mode GaN with a silicon driver in a cascode configuration.

An alternative approach is to redesign the device as normally-off enhancement mode.  Enhancement-mode devices have other issues, such as different driver requirements than an equivalent silicon device. For example, the maximum allowable V_GS for an enhanced GaN FET is 6 V—lower than for silicon.

For both types, the higher switching frequency of a GaN transistor demands greater timing precision in the switch-driving signal. GaN switches are also highly sensitive to parasitic impedances from packages, interconnects, and outside sources.

Manufacturers offer both general-purpose GaN drivers and integrated solutions for defined applications. TI’s UCC28780, for example, is an ACF controller with ZVS capability that offers switchable GaN or Si drive circuits. The ZVS portion incorporates auto-tuning, adaptive dead-time optimization, a variable switching-frequency control law, and adaptive multimode control that changes operation based on input and output conditions. These features allow the UCC28780 to achieve ZVS over a wide operating range.

Integrating a GaN FET, a driver, and protection circuitry into a single package is another option. The LMG341xR070 (Fig. 10) provides an alternative to cascode GaNs or standalone devices with separate drivers. The device’s gate driver enables 100-V/ns switching with near-zero V_DS ringing; current-limiting circuitry protects against unintended shoot-through events. Overtemperature shutdown prevents thermal runaway, and system interface signals provide self-monitoring capability.

10. The LMG341xR070 combines a 600-V GaN FET with a driver and protection circuitry. (Source: TI Products: LMG3410R0710)

Resources

Texas Instruments has multiple resources to help engineers add GaN devices in their designs. The GaN overview page is a good place to start; it contains GaN-related white papers, application notes, product pages, and reference designs.

We discussed the operation of the ACF in this April 2018 article. Pei-Hsin Liu’s paper at the 2018 Texas Instruments Power Supply Design Seminar also reviews ACF operation in great detail.  The seminar includes a set of training videos on the same topics, too. For another example of GaN-related improvements, consult this series of videos on resonant converter topologies.

 Sponsored Resources: 

• Power Supply Design Seminar Training Series
• Survey of Resonant Converter Topologies
• Comparison of GaN and Silicon FET-Based Active Clamp Flyback Converters